The usage of light-emitting diodes (LEDs) to provide illumination is increasing rapidly as the cost of LEDs decrease and the endurance of the LEDs increases to cause the overall effective cost of operating LED lighting products to be lower than incandescent lamps and fluorescent lamps providing equivalent illumination. Also, LEDs can be dimmed by controlling the current through the LEDs because LEDs are current driven devices. The current through a plurality of LEDs in a lighting device must be controlled tightly in order to control the illumination provided by the LEDs. Typically, the secondary of an LED lighting device must be electrically isolated from the primary (line and neutral side) of the lighting device to meet applicable safety standards (e.g., IEC class II isolation). In addition, an LED driver circuit should have a high power factor and should have a constant current control.
One known solution to the foregoing requirements is to use a flyback converter to produce the DC in the secondary from the primary source. The flyback converter provides power factor correction to produce a high power factor, provides isolation between the primary and secondary circuits, and has a reasonably low cost. By using primary current sensing techniques to control the secondary current through the LEDs, the flyback converter provides an LED driver that is low in cost when compared with other topologies.
FIG. 1 illustrates a typical LED driver circuit 100 based on a flyback converter. The LED driver circuit provides current to an LED load 110. In the illustrated embodiment, the LED load may comprise from two to five LEDs 112 connected in series between a first (+) LED load terminal 114 and a second (−) LED load terminal 116. A common load current flows through each LED in the LED load to cause the LEDs to illuminate. In alternative embodiments, the LED load may comprise additional LEDs in series or a series-parallel combination of LEDs. In order to provide consistent illumination, the load current through the LEDs should be maintained at a substantially constant magnitude.
The driver circuit utilizes a primary current sensing technique to control the secondary current. In the driver circuit, an AC source 120 provides an AC input voltage via first AC input line 122 and a second AC input line 124. In the illustrated embodiment, the AC input voltage may vary from 86 volts RMS to 265 volts RMS.
The AC input voltage between the first AC input line 122 and the second AC input line 124 is applied between a first input terminal 132 and a second input terminal 134 of a full-wave bridge rectifier 130. The bridge rectifier has a first (+) output terminal 136 and a second (−) output terminal 138. A first rectifier diode 140 has an anode connected to the first input terminal and a cathode connected to the first output terminal. A second rectifier diode 142 has an anode connected to the second input terminal and a cathode connected to the first output terminal. A third rectifier diode 144 has an anode connected to the second output terminal and has a cathode connected to the first input terminal. A fourth rectifier diode 146 has an anode connected to the second output terminal and has a cathode connected to the second input terminal. The bridge rectifier operates in a conventional manner to produce a pulsating DC voltage on the first output terminal which is referenced to the second output terminal. The second output terminal is connected to a primary ground reference 150.
The first (+) output terminal 136 of the bridge rectifier 130 is connected to a first terminal 164 of the primary winding 162 of an isolation transformer 160. The primary winding of the isolation transformer has a second terminal 166. The isolation transformer has a secondary winding 170, which has a first terminal 172 and a second terminal 174. The isolation transformer has an N:1 turns ratio between the primary winding and the secondary winding such that the voltage across the primary winding is N times the voltage across the secondary winding and such that the current through the secondary winding is N times the current through the primary winding.
As further illustrated in FIG. 1, the first terminal 172 of the secondary winding 170 of the isolation transformer 160 is connected to secondary ground reference 180. The secondary ground reference is electrically isolated from the primary ground reference 150. The second terminal 174 of the secondary winding is connected to the anode of a secondary diode 182. The cathode of the secondary diode is connected to the first (+) terminal 186 of a secondary filter capacitor 184. The secondary filter capacitor may also be referred to as an output filter capacitor. A second (−) terminal 188 of the secondary filter capacitor is connected to the secondary ground reference and thus to the first terminal of the secondary winding of the isolation transformer. In one embodiment, the secondary filter capacitor has a capacitance of approximately 2,000 microfarads. The cathode of the secondary diode and the first terminal of the secondary filter capacitor are connected to a first (+) output terminal 190 of the LED driver circuit 100. The secondary ground reference is connected to a second (−) output terminal 192 of the LED driver circuit.
As illustrated in FIG. 1, dots on the terminals of the primary winding 160 and the secondary winding 170 of the isolation transformer represent the magnetic coupling between the two windings. When the first terminal 162 of the primary winding is positive with respect to the second terminal 164 of the primary winding, the first terminal 172 of the secondary winding is also positive with respect to the second terminal 174 of the secondary winding; however, the current flow through the secondary winding is opposite the current flow through the primary winding. Thus, when current flows into the first terminal of the primary winding and flows to the second terminal of the primary winding with an increasing magnitude, the increasing current flow should induce current to flow from the second terminal of the secondary winding to the first terminal of the secondary winding (e.g., downward through the secondary winding toward the secondary ground reference 180) when the secondary winding terminals are oriented as shown in FIG. 1). However, induced current flow in that direction is blocked by the reverse-biased secondary diode 182. In contrast, when the magnitude of the current flowing from the first terminal to the second terminal of the primary winding decreases, current flow is induced in the secondary winding that flows from the first terminal of the secondary winding to the second terminal of the secondary winding (e.g., upward through the secondary winding when the secondary winding terminals are oriented as shown in FIG. 1). The current flowing out of the second terminal of the secondary winding passes through the forward-biased secondary diode.
The first (+) output terminal 190 of the LED driver circuit 100 is connected to the first (+) terminal 114 of the LED load 110. The second (−) terminal 116 of the LED load is connected to the secondary ground reference 180 via the second (−) output terminal 192 of the LED driver circuit.
As further shown in FIG. 1, the second terminal 166 of the primary winding 162 of the isolation transformer 160 is connected to a first terminal 202 of a semiconductor switch 200. The switch further includes a second terminal 204 and a control terminal 206. For example, the semiconductor switch may comprise a metal oxide semiconductor field effect transistor (MOSFET) wherein the first terminal is the drain of the MOSFET, the second terminal is the source of the MOSFET, and the control terminal is the gate of the MOSFET. In the illustrated embodiment, the MOSFET is an N-channel enhancement mode transistor, which has is normally off (e.g., has a high impedance between the drain and the source). The MOSFET turns on to provide a low-impedance path (e.g., a few tens of milliohms) between the drain and the source when a sufficiently large voltage differential is applied between the gate and the source of the MOSFET.
The second terminal (source) 204 of the MOSFET 200 is connected to a primary sensing node 210. The primary sensing node is also connected to a first terminal 214 of a primary sensing resistor 212. A second terminal 216 of the primary sensing resistor is connected to the primary ground reference 150. When the MOSFET is turned on, a current flows from the first (+) output terminal 136 of the bridge rectifier 130, through the primary winding 162 of the isolation transformer 160, through the MOSFET from the first terminal (drain) 202 to the second terminal (source), and through the primary sensing resistor to the primary ground reference.
The control terminal (gate) 206 of the MOSFET 200 is controlled by a gate drive (GD) output terminal 222 of an LED lighting controller integrated circuit (“control IC”) 220. The LED lighting controller receives a feedback voltage via a current sense (CS) input terminal 224, which is connected to the primary sensing node 210. Thus, the LED controller receives a voltage proportional to the instantaneous current flowing through the primary sensing resistor 212. The current flowing through the primary sensing resistor is the same current flowing through primary winding 162 of the isolation transformer 160. Thus, the voltage developed across the primary sensing resistor from the primary sensing node to the primary ground reference 150 is proportional to the current flowing through the primary winding of the isolation transformer. Accordingly, the current sense input terminal of the LED lighting controller receives a voltage proportional to the current flowing through the primary winding of the isolation transformer.
In the illustrated embodiment, the LED lighting controller 220 comprises an MP4027 primary-side-control, offline LED lighting controller commercially available from Monolithic Power Systems (MPS) of San Jose, Calif. The LED lighting controller includes additional inputs (e.g., power input, ground reference, and compensation inputs), which are not shown in FIG. 1.
The LED lighting controller 220 operates in a conventional manner to output a high output signal on the gate control output terminal 222 to turn on the MOSFET 200 to cause current to flow through the primary winding 162 of the isolation transformer 160 from the first terminal 164 to the second terminal 166 of the primary winding. The current is allowed to build up to a selected magnitude in the primary winding as sensed via the primary sensing resistor 212. As discussed above, no current flows in the secondary winding 170 of the isolation transformer while current is flowing in the primary winding because the secondary diode 182 connected to the second terminal 174 of the secondary winding is reversed biased.
When the current flowing in the primary winding 160 reaches the selected magnitude, the LED lighting controller 220 turns off the MOSFET 200. The energy stored in the isolation transformer 160 is transferred to the secondary winding 170 and current flows out of the second terminal 174 of the secondary winding through the secondary diode 182 to the secondary filter capacitor 184 to charge the secondary filter capacitor. The voltage built up on the secondary filter capacitor is applied across the LED load 110 to provide current to cause the LEDs 112 within the LED load to illuminate.
The LED lighting controller 220 operates in a manner described in detail in the data sheets provided by the supplier to produce a DC current IOUT through the LED load 110 that is determined by the value of the primary sensing resistor 212. As set forth in the data sheets for the MP4027 controller from Monolithic Power Systems, the current through the LED load is controlled in accordance with the following approximate relationship:IOUT≈(N×VREF)/(2×RSENSE)                where        IOUT is the current flowing through the LED load;        N is number of primary turns per each secondary turn;        VREF is an internal reference voltage within the LED lighting controller; and        RSENSE is the resistance of the primary sensing resistor.        
In one embodiment, the internal reference voltage VREF of the LED lighting controller 220 is approximately 0.413 volts, N is 5, and RSENSE is approximately 5.72 ohms. Thus, LED lighting controller controls the current through the LED load 110 to be approximately 180 milliamperes.
The LED driver circuit 100 shown in FIG. 1, which uses a flyback converter with primary current sensing, provides a simple and cost effective way to control the current through the LED load 110; however, testing has shown that the circuit in FIG. 1 does not provide adequate current regulation over a wide range of input voltages and over a range of the number of LEDs 112 in series within in the LED load. Furthermore, the ripple in the output current through the LED load is excessive.
Table 1 illustrates the results of testing the circuit of FIG. 1 over a range of input voltages and over a range of the number of LEDs connected in series within the LED load 110.
TABLE 1Flyback Primary Sensing LED Driver Testing Data with 2,000 μF Output CapacitorLineLoadNo. ofVINPINIOUT_AVGIOUT_MAXIOUT_MINVOUTRippleReg.Reg.LEDs(volts)(watts)(mA)(mA)(mA)(volts)(%)(%)(%)5 863.46179.720915013.8616.3015.752653.87181.520715313.8614.051.004 862.8418522014911.118.922653.318922115311.116.932.163 862.231902351458.423.682652.721982391518.420.714.212 861.61962561335.630.612652.12082651435.627.406.12
As illustrated in Table 1, the target LED load current is 180 milliamperes. The input voltages from the AC source 120 range from approximately 86 volts RMS to approximately 265 volts RMS. The LED load 110 ranges from two LEDs 112 in series (requiring approximately 5.6 volts across the LED load) to five LEDs in series (requiring approximately 13.9 volts across the LED load). The testing data in Table 1 show that the line regulation (the change in average LED current caused by input line voltage change) can be as high as 6.12% for two LEDs in series within the LED load. The testing data in Table 1 further show that the load regulation (the change in LED current in response to changes in the LED load over the input voltage range) is approximately 15.75% (e.g., [(208 mA−179.7 mA)/179.7 mA]×100=15.75%). As further illustrated in Table 1, the ripple of the current through the LED load is measured to be in a range from approximately 14% to approximately 31% over a range of input voltages and a range of LEDs connected in series.
For acceptable operation, an LED driver circuit 100 should provide line regulation and load regulation of the current to be less than 5% and should maintain the ripple to be less than 20%. Although the ripple can be reduced by increasing the capacitance of the secondary filter capacitor 184, filter capacitors having capacitances greater than 2,000 microfarads at the required voltages can be quite large and expensive. The size and cost at least partially reduce the cost effectiveness and practicality of the illustrated circuit.
The primary side current sensing technique performed by the LED driver circuit 100 of FIG. 1 is based on the above-described approximate relationship between the measured voltage across the primary sensing resistor 212 on the primary side and the actual current through the LED load 110 on the secondary side. Because of the approximate relationship, the LED current regulation cannot be improved easily by modifying the primary side of the driver circuit.
Accordingly, a need exists for a circuit and a method for improving the current regulation and reducing the output current ripple of the LED driver circuit based on a flyback converter that is controlled by primary side current sensing.